Power Supply Design

Designing a power supply well requires understanding what the load needs (voltage accuracy, transient response, noise limits, isolation), what the input looks like (AC mains, battery, regulated bus), and how to connect them efficiently and safely. The protection circuits matter as much as the regulation itself — an unprotected supply can fail catastrophically and take the load with it.

Design Checklist

Requirement	How to specify it	Affects
Output voltage	Nominal ± tolerance (e.g., 5V ±2%)	Regulator topology, feedback accuracy
Output current	Maximum continuous + peak surge	Transistor ratings, transformer/inductor sizing
Input voltage range	Min and max (battery sag + overcharge, mains variation)	Dropout headroom, duty cycle limits
Efficiency	Required % at typical load	Linear vs switching; switching frequency
Output noise	mVpp ripple at specific frequency, or µV RMS in bandwidth	Topology choice; may require linear post-regulator after switcher
Transient response	Max voltage deviation at max dI/dt step	Control loop bandwidth, output capacitance
Isolation	Required or not; withstand voltage	Transformer-based topology mandatory if yes
Operating temperature	Min/max ambient	Component derating; heatsink and fan requirements

Linear Supply Design — Transformer to Output

The progression of power from AC mains to regulated DC. The transformer steps down the voltage; the rectifier folds the negative halves positive; the filter capacitor smooths the bumps into a continuous DC with ripple; and the linear regulator burns off the excess to provide a perfectly flat output.

Step 1 — Transformer Secondary Voltage

After full-wave rectification and filtering, the DC voltage is approximately:
  V_DC = V_secondary_RMS × 1.414 − 2 × V_diode_forward

Needed V_DC:
  V_DC_min = V_out + V_dropout + V_ripple_peak

Example: 12V output, 78xx (V_dropout=2V), 2V ripple budget
  V_DC_min = 12 + 2 + 2 = 16V
  V_secondary = 16V / 1.414 ≈ 11.3V → use 12V RMS transformer

Transformer VA rating:
  VA = V_secondary × I_secondary × 1.5 (derating factor for rectifier current spikes)
  At 1A output: VA = 12V × 1A × 1.5 = 18VA → use 20 or 25VA transformer

Step 2 — Filter Capacitor

V_ripple = I_load / (f_ripple × C)

  f_ripple = 2 × f_mains (120Hz for NA, 100Hz for EU/UK — full-wave)

Solving for C:
  C = I_load / (f_ripple × V_ripple_allowed)

Example: 1A load, 120Hz, 2V ripple budget:
  C = 1 / (120 × 2) ≈ 4167µF → use 4700µF

For 2A, same budget:
  C = 2 / (120 × 2) ≈ 8333µF → use 10,000µF

Capacitor voltage rating: V_rating ≥ 1.5 × V_peak
  V_peak from 12V RMS transformer: 12 × 1.414 = 17V
  Use 25V or 35V rated electrolytic capacitors

Step 3 — Regulator and Dissipation

Power dissipated = (V_in − V_out) × I_load

At V_in=16V, V_out=12V, I_load=1A:
  P = (16-12) × 1 = 4W → heatsink required

Heatsink requirement:
  T_junction = T_ambient + P × θ_JA
  For 78xx TO-220: θ_JC = 5°C/W
  Required θ_total ≤ (125 − 25 − 20°C safety) / P = 80 / 4 = 20°C/W
  θ_JC = 5°C/W, θ_CS ≈ 0.5°C/W
  Required heatsink: θ_SA ≤ 14.5°C/W

Switching Supply Design — Buck Converter

During the ON phase, energy is drawn from the input and stored in the inductor's magnetic field. During the OFF phase, the inductor acts as the source, pumping its stored energy into the load through the flyback diode.

Parameter	Formula	Example (12V→5V, 2A, 300kHz)
Duty cycle D	D = V_out / V_in (ideally)	D = 5/12 = 0.417
Inductor ripple current ΔI_L	Typically 20–40% of I_out	ΔI_L = 0.6A (30% of 2A)
Inductance L	L = (V_in − V_out) × D / (f_sw × ΔI_L)	L = 7 × 0.417 / (300k × 0.6) = 16.3µH → 22µH
Inductor peak current	I_peak = I_out + ΔI_L/2	I_peak = 2 + 0.3 = 2.3A → rate inductor at ≥3A Isat
Output capacitor C_out	C = ΔI_L / (8 × f_sw × V_ripple)	C = 0.6 / (8 × 300k × 0.05V) = 500µF → 470µF low-ESR
Input capacitor C_in	C_in = I_out × D × (1−D) / (f_sw × V_ripple_in)	C_in = 2 × 0.417 × 0.583 / (300k × 0.1) ≈ 16µF → 22µF

Switching Frequency Trade-offs

Frequency	Inductor size	Cap size	Efficiency	EMI
50–100 kHz	Large (100µH+)	Larger	Higher (lower switching loss)	Lower (easier to filter)
200–500 kHz	Medium (10–47µH)	Medium	Good balance	Moderate
1–4 MHz	Small (1–10µH)	Small ceramic	Lower (switching losses dominate)	Higher frequency, easier filter but harder shielding

EMI and Filtering

Switching supplies are inherently noisy. The fast switching edges contain high-frequency harmonics that radiate and conduct back into the mains:

Critical layout for minimum EMI (buck converter):
  Switching loop: FET → inductor → output cap → back to FET source
  Keep this loop SMALL — it's the primary radiating antenna
  Minimise loop area by placing FET, diode, and input cap tightly

Input EMI filter (before the converter):
  Common-mode choke: blocks CM noise returning to mains
  X-capacitors: across the supply line (line-to-line)
  Y-capacitors: from each line to chassis (line-to-earth)
  Minimum: ferrite bead + 100nF ceramic on input
  For conducted emissions compliance: proper LC input filter

Output ripple suppression:
  Main output cap + small LC filter (1µH + 10µF ceramic):
    Attenuates switching ripple by additional 40dB at f_sw
    Critical for noise-sensitive analog loads

Protection Circuits

Overvoltage Protection (Crowbar)

Crowbar circuit:
  Zener (Vz = V_out_max) in series with SCR gate
  SCR anode to output, cathode to GND (via fuse in series)

  Normal: V_out < Vz → SCR off
  Fault:  V_out > Vz → Zener conducts → SCR fires → shorts output → fuse blows

  Latching protection: SCR stays on until power cycled (fuse must blow first)

TL431-based precision crowbar:
  Reference voltage set by resistor divider
  ±0.5% trip accuracy vs ±5% for simple Zener

Overcurrent / Short Circuit Protection

Current sense resistor + comparator:
  R_sense in series with output (0.05–0.1Ω)
  V_sense = I_out × R_sense
  At I_max: V_sense = I_max × R_sense > V_threshold → comparator trips → enable pin low

Foldback current limiting (safer for linear regulators):
  Above I_limit: output voltage folds back as current increases
  At short circuit: output voltage ≈ 0, current << I_limit
  Reduces power dissipation during fault (vs constant-current limiting)

Component: LM317 has built-in 1.5A current limiting and thermal shutdown.
  No external sense resistor needed for basic protection.

Reverse Polarity Protection

Method	Voltage drop (forward)	Protection	Cost
Series diode (1N5817 Schottky)	0.3–0.5V (always)	Good	Lowest
P-MOSFET series (gate to output)	I × RDS_on ≈ 1–50mV	Good, lower drop	Low
Fuse + diode in parallel	0V (forward), fuse blows reverse	Good (destructive)	Low
Dedicated protection IC (e.g., MAX16054)	Very low	Excellent — includes transient protection	Moderate

The P-MOSFET conducts when forward-biased because Vgs < 0. When reverse-biased, Vgs > 0 and it blocks current. (Note: standard designs often tie the gate to ground rather than the output rail).

Forward: Vgs = V_out − V_in < 0 → MOSFET on → passes power
Reverse: Vgs = V_out − V_in > 0 → MOSFET off → blocked

Testing Checklist

Test	What to measure	Accept if
No-load output voltage	V_out at zero current	Within ±1% of nominal
Full-load output voltage	V_out at I_max	Load regulation within spec
Output ripple	AC voltage on output (scope, AC coupled, 20MHz BW)	Less than specified (mVpp)
Transient response	V_out deviation on sudden load step (0→50%→100%→50%→0)	Settles within spec time, overshoot within limits
Efficiency	(V_out × I_out) / (V_in × I_in) × 100%	Meets spec at typical load
Short circuit	Short output, measure supply current, check no damage	Current limits to rated maximum; recovers on removal of short
Thermal	Infrared thermometer or thermocouple at full load, 30 min	All components below derating threshold
Input range	V_out at V_in_min and V_in_max	Within regulation spec across full input range

References

Erickson & Maksimovic — Fundamentals of Power Electronics, 3rd ed.
Mohan, Undeland & Robbins — Power Electronics, 3rd ed.
Maniktala — Switching Power Supply Design and Optimization
TI WEBENCH — complete switching supply calculator and simulation ↗
Linear Technology AN19 — LT1070 Design Manual (PDF) — classic switcher design reference ↗